1. Field of the Invention
This invention relates to the field of analog circuits, and, in particular, to continuous-time filters.
2. Background Art
Seven pole electronic filters with variable high frequency boost are widely used in zoned bit signal processing channels of hard disk drives to optimize the cutoff frequency (Fc) and boost for variable data transfer rates. The high frequency boost serves to slim incoming data pulses to reduce intersymbol interference.
Typical seven pole filters, such as that shown in FIG. 1, consist of three biquadratic filter sections, or "biquads," followed by a single order output section. A feedforward section in front of the first biquad provides the pulse-slimming function. An example of a biquad element with feedforward boost is discussed with respect to FIG. 9 below.
In FIG. 1, input signal IN 100 is provided to the filter input of the first biquad element, BIQ1 102. The filter input stage of the biquad is commonly a transconductance stage, represented by block 110 labeled "Gm" in FIG. 1. Input signal IN 100 is also provided to feedforward amplifier 101 to provide a programmable gain ALPHA for feedforward signal 109.
Feedforward signal 109 is provided to BIQ1 102 where signal 109 is AC coupled to a summing node to provide high frequency boost to the filter signal containing the input signal 100 component. The output of BIQ1 102 is coupled to the input of the second biquad element, BIQ2 103, and the output of BIQ2 103 is coupled to the input of the third biquad element, BIQ3 104. The output of BIQ3 104 is coupled to single-pole lowpass output section 105 and single-pole highpass output section 106. Standard or "normal" filter output 107 is provided by section 105. Differentiated filter output 108 is provided by section 106.
Each "BIQ" block in FIG. 1 is a 2nd order "Biquad" circuit with electronically variable cutoff frequency Fc controlled by a digital-to-analog converter (DAC). In the normalized transfer function of the biquad circuit, the cutoff frequency is assumed to be 1 radian/second or 1/2.pi. hertz. This function can be denormalized for actual cutoff frequencies by replacing the complex frequency variable "s" with s/2.pi.Fc) where Fc is in hertz. The high frequency boost is increased by increasing the boost value ALPHA in the feedforward section, which is typically in the range between zero and five.
FIG. 2 is a block diagram showing the transfer functions of the respective sections of the seven pole filter. Sections 101 and 102 are combined into Boost Adjust section 200 with a biquadratic transfer function of: ##EQU1## where .omega..sub.1 and Q.sub.1 are the cutoff frequency and quality factor of the first biquad BIQ1 102. ALPHA is the gain of the feedforward amplifier 101. Biquad sections 103 and 104 provide the transfer functions of: ##EQU2## where k=2 for BIQ2 and k=3 for BIQ3. Section 105 provides a lowpass transfer function of: ##EQU3## and section 106 provides a highpass transfer function of: ##EQU4## The transfer function provided by the entire seven pole filter is obtained by multiplying the transfer functions of equations 1-3, or 1-2 and 4, as appropriate for the filter sections involved, into a single seventh-order expression.
FIG. 9 illustrates one biquad implementation including a feedforward boost function. Other biquad circuits exist, such as non-differential and/or resistive RC implementations. FIG. 9 is a circuit block diagram of a differential biquad circuit of transconductance amplifiers and capacitors. The cutoff frequency of the circuit is determined by the gm/C values within the main feedback loop (elements C1-C4, 904 and 907). Element 101 is the feedforward amplifier with gain of .alpha. (ALPHA). The "Gm" element 110 in biquad 102 of FIG. 1 corresponds to transconductance amplifier 902. The AC coupling capacitor in biquad 102 corresponds to capacitors C3 and C4.
Input signal IN 100, in differential form, is provided to the input ports of transconductance amplifier 902 and feedforward amplifier 101. The positive and negative output ports of transconductance amplifier 902 are coupled through capacitors C1 and C2, respectively, to stable reference voltage 900. Further, the positive and negative output ports of transconductance amplifier 902 are coupled to the positive and negative input ports of buffer 903, as well as the negative and positive output ports, respectively, of transconductance amplifier 907. The positive and negative output ports of buffer 903 are coupled to the positive and negative input ports of transconductance amplifier 904. Buffers 903 and 905 are utilized for level-shifting and biasing purposes.
The positive and negative output ports of feedforward amplifier 101 provide feedforward signal 109. The positive output port of feedforward amplifier 101 is coupled through capacitor C4 to the negative input port of buffer 905. The negative output port of transconductance amplifier 904 and the positive output port of transconductance amplifier 906 are also coupled to the negative input port of buffer 905. The negative output port of feedforward amplifier 101 is coupled through capacitor C3 to the positive input port of buffer 905.
The positive output port of transconductance amplifier 904 and the negative output port of transconductance amplifier 906 are also coupled to the positive input port of buffer 905. The input port of buffer 905 is the summing node where AC coupled feedforward signal 109 is internally summed with the filter signal. The positive and negative output ports of buffer 905 are coupled to the positive and negative input ports of transconductance amplifiers 906 and 907. The lowpass output of the biquad is taken across the positive and negative output ports of buffer 905 (marked as Vlp+ and Vlp-).
If capacitors C1-C4 are chosen to have the same value, the frequency response of the biquad of FIG. 9 is given by the following equation: ##EQU5## The transconductance values of transconductance amplifiers 902, 903, 906 and 907 are gm1, gm3, gm2 and gm1, respectively.
A disadvantage of the electronic filters of the prior art is that as boost (ALPHA) is increased, the absolute gain of the filter for all frequencies beyond DC, and especially higher frequencies in the vicinity of the cutoff frequency, increases significantly. Rapid boost changes typically occur for electronic filters used in read/write channels of sector servo disk drives when changing a read/write channel from "data" mode to "servo" mode. The increase in gain must be compensated by an automatic gain control loop (AGC) as well as any DC offset restoring circuits in the channel. The result is an increase in overhead time and disk space required for the transients to settle out. Disk drive performance is degraded due to the limitations imposed on data storage density and data access time. Furthermore, the gain shift artificially consumes a large portion of the dynamic range of the AGC that could otherwise be used to regulate additional head/disk signal variations.
FIG. 3 shows a family of boost magnitude response curves with Fc of 22 MHz. For this figure, the components of the normalized transfer function are: ##EQU6## Curves are plotted for ALPHA=0, 1, 3 and 5. For the normalized curves, the DC value is 0 dB. As shown, for increasing ALPHA, the high frequency gain rises rapidly above 0 dB. For ALPHA=5, the gain boost reaches 13 dB. While compensating for intersymbol interference, the large increase in high frequency gain places unwanted demands on the AGC. Thus, in the prior art, the AGC must account for the large gain shift during events such as the switching of a read channel between "data" and "servo" modes.